Power transistor driving circuits and methods for switching mode power supplies

ABSTRACT

A power supply controller is provided for providing a drive current to a control terminal of a power transistor in three time intervals. The controller includes control circuits configured to control the drive current in multiple stages. During a first time interval, first drive current includes a current spike for turning on the power transistor in response to a start of the control signal pulse. During a second time interval, a second drive current includes a ramping current substantially proportional to a magnitude of a current through the power transistor. During a third time interval, current flow to the power transistor is at least partially turned off before an end of the control signal pulse.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.12/638,809 filed Dec. 15, 2009, which claims priority to Chinese PatentApplication No. 200910226104.3, filed Nov. 20, 2009 by inventors JianhuaDuan et. al., commonly assigned and incorporated in their entirety byreference herein for all purposes.

BACKGROUND OF THE INVENTION

Embodiments of the present invention are directed to power supplycontrol circuits and power supply systems. More particularly,embodiments of the present invention provides methods and circuits forcontrolling a power transistor in a switching mode power supply (SMPS).

Regulated power supplies are indispensable in modern electronics. Forexample, the power supply in a personal computer often needs to receivepower input from various outlets. Desktop and laptop computers oftenhave regulated power supplies on the motherboard to supply power to theCPU, memories, and periphery circuitry. Regulated power supplies arealso used in a wide variety of applications, such as home appliances,automobiles, and portable chargers for mobile electronic devices, etc.

In general, a power supply can be regulated using a linear regulator ora switching mode controller. A linear regulator maintains the desiredoutput voltage by dissipating excess power. In contrast, a switchingmode controller rapidly switches a power transistor on and off with avariable duty cycle or variable frequency and provides an average outputthat is the desired output voltage.

Compared with linear regulators, switching mode power supplies have theadvantages of smaller size, higher efficiency and larger output powercapability. On the other hand, they also have the disadvantages ofgreater noise, especially Electromagnetic Interference at the powertransistor's switching frequency or its harmonics.

Pulse Width Modulation (PWM) and Pulse Frequency Modulation (PFM) aretwo control architectures of switching mode power supplies. In recentyears, green power supplies are emphasized, which require higherconversion efficiency and lower standby power consumption. In a PWMcontrolled switching mode power supply, the system can be forced toenter into burst mode in standby conditions to reduce power consumption.In a PFM controlled switching mode power supply, the switching frequencycan be reduced in light load conditions. PFM-controlled switching modepower supply exhibits simple control topology and small quiescentcurrent. Therefore, it is suitable for low cost small output powerapplications such as battery chargers and adapters.

In such a switched mode power supply system, a switch is connected tothe primary winding of the transformer. In the switching power supplies,the power transistor switch on and off periodically to convert theprimary current of the transformer to the secondary side. The stableoutput voltage will be obtained by regulating the duty cycle orfrequency of the primary side switch. Magnetic energy is stored in theinductance of the primary winding when the switch is turned on, and theenergy is transferred to the secondary winding when the switch is turnedoff. The energy transfer results in a current flowing through thesecondary winding and the rectifying diode. When the energy transfer iscompleted, i.e., the current stops flowing through the diode, asubstantially sinusoidal oscillation of decreasing amplitude appears atthe secondary winding.

The frequency of the sinusoidal oscillation is determined, in part, bythe inductance of the primary winding and by the parasitic capacitancein the primary winding as well in the printed circuit board. The effectof these components often are difficulty to determine in advance and canlead to output performance limitations in the switched mode powersupply. Some of these limitations are described in more detail below.

Therefore, there is a need for techniques that can provide moreeffective control of the power transistor in a switching mode powersupply.

BRIEF SUMMARY OF THE INVENTION

Embodiments of the present invention provides methods and circuits forcontrolling a power transistor in a switching mode power supply (SMPS).In a specific embodiment, a method and circuits is provided formulti-stage turn on and turn off the power transistor to achieve reducedswitching loss and better conversion efficiency. But it would berecognized that the invention has a much broader range of applicability.

In an embodiment, a power supply controller is provided for providing adrive current to a control terminal of a power transistor in three timeintervals. The controller includes control circuits configured tocontrol the drive current in multiple stages:

-   -   during a first time interval, provide a first drive current that        includes a current spike for turning on the power transistor in        response to a start of the control signal pulse;    -   during a second time interval, provide a second drive current        that includes a ramping current substantially proportional to a        magnitude of a current through the power transistor; and    -   during a third time interval, at least partially turn off        current flow to the power transistor before an end of the        control signal pulse.

In some embodiments, the power transistor is first turned on by aninitial overdrive current, and the overdrive current is furtherincreased by mirroring an leading edge current spike in a feedbacksignal representing the primary winding current. After the terminationof the overdrive stage, the drive current is increased proportionallywith the increase of the primary winding current during the primarywinding on time. When the primary winding current reaches apre-determined level, the main driving current is cutoff for a timeperiod. In an embodiment, this cutoff time period is selected based on astorage time of the power transistor.

In some embodiments, when it is time to cut off the remaining forwarddriving current, the power transistor turn off stage starts with thedischarge of storage charges rapidly by overdriving a dischargetransistor coupled to the power transistor. And after the initialdischarge of the storage charge, a moderate Rdson of the dischargetransistor is maintained. In this way, a large reverse current isgenerated to provide a rapid turn-off of the power transistor. As aresult, lower switching loss is achieved. More details of theseembodiments are described below in the context of a bipolar powertransistor.

In another embodiment, a method is provided for short circuit protectionof the power supply, which could be damaged if the current sensingcircuit on the primary side malfunctions. For example, if the currentsense resistor is shorted or is changed to a lower value, a largecurrent may flow in the power supply circuit. In embodiments of theinvention, with the multistage turn-on methods, the power transistor isonly turned on by the initial overdrive current. And virtually nofurther base drive current is generated due to the lower voltage on thecurrent sense resistor, so the power transistor will turn on for a shortperiod of time and then be cut off. In this embodiment, the primarycurrent is prevented from continuing to increase indefinitely, thusensuring the safety of the power transistor and the whole system. Thismethod is more reliable and efficient compared with conventional drivingmethods.

Another embodiment provides a driver circuit for turning off a bipolartransistor. The driver circuit includes a current mirror having firstand second PMOS transistors coupled to a first voltage during a firsttime period, a bipolar transistor coupled to the current mirror, and aplurality of diode devices coupled between a collector and a base of thebipolar transistor. The driver circuit also has an MOS transistor havinga gate terminal coupled to the collector of the bipolar transistor and adrain terminal for coupling to a control terminal of the bipolartransistor for turning off the power transistor.

Another embodiment provides a method for driving a bipolar powertransistor. The method includes, during a first time period, turning onthe power transistor with an initial drive current, and increasing thedrive current with a current related to a leading edge current spike ona current sense resistor, and maintaining current flow in the powertransistor with a ramping drive current. The method also includes,during a second time period, turning off the drive current whileallowing current flow in the power transistor to continue.

According to another embodiment, a method for turning off a bipolartransistor includes coupling an MOS transistor to a base of the bipolartransistor. During a first time period, the method includes applying afirst voltage to a gate of the MOS transistor to discharge the bipolartransistor. During a second time period, the method includes applying asecond voltage to the gate of the MOS transistor, the second voltagebeing lower than the first voltage.

Another embodiment provides a switching mode power supply (SMPS) thatincludes a primary winding, a power transistor coupled to the primarywinding, and an auxiliary winding for providing a feedback signal, whichis related to a current flow through the primary winding. The SMPS alsohas a power supply controller for providing a drive current to a controlterminal of a power transistor in three time intervals. Depending on theembodiment, the controller can have one or more of the various featuresdescribed above.

These and other features and advantages of embodiments of the presentinvention will be more fully understood and appreciated uponconsideration of the detailed description of the preferredimplementations of the embodiments, in conjunction with the followingdrawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram illustrating a conventional switching modepower supply system;

FIG. 2 is a timing diagram illustrating signal waveforms in aconventional switching mode power supply system;

FIG. 3A is a circuit schematic/block diagram illustrating a switchingmode power supply system according to an embodiment of the presentinvention;

FIG. 3B is a timing diagram illustrating signal waveforms in the powersupply controller of FIG. 3A according to an embodiment of the presentinvention;

FIG. 3C is a circuit schematic/block diagram illustrating a switchingmode power supply system including specific implementation of the drivercircuit of FIG. 3A according to an embodiment of the present invention;

FIG. 3D is a timing diagram illustrating waveforms of various signals inthe switching mode power supply system of FIG. 3C according to anembodiment of the present invention;

FIG. 4 is a circuit diagram illustrating a specific implementation ofthe bias block circuit of the switching mode power supply system of FIG.3C according to an embodiment of the present invention;

FIG. 5A is a circuit diagram illustrating a specific implementation ofthe pre-shutdown circuit of the switching mode power supply system ofFIG. 3C according to an embodiment of the present invention;

FIG. 5B is a timing diagram illustrating waveforms of various signals inthe pre-shutdown circuit of FIG. 5A according to an embodiment of thepresent invention; and

FIG. 6 is a circuit diagram illustrating a specific implementation ofthe overdrive block circuit the switching mode power supply system ofFIG. 3C according to an embodiment of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 is a schematic diagram illustrating a conventional switching modepower supply system (SMPS) 100. As shown, SMPS 100 is a flyback-typeprimary side regulated switched mode power supply using a pulsefrequency modulation (PFM) control. The primary side regulated switchedmode power supply includes a rectifier and capacitor (not shown) thatconverts an AC mains voltage to an unregulated voltage Vin, which isthen applied to a terminal of a primary winding 114 of a transformer102. Primary winding 114 is also coupled to a high voltage NPN powertransistor 101, which is turned on and off by a signal OUT of a powersupply controller 103. When transistor 101 is turned on, a primarycurrent Ip flows through primary winding 114, which starts to build up amagnetic energy. A secondary winding 115 is magnetically coupled toprimary winding 114. When transistor 101 is turned on, the magneticfield produced by a primary current Ip induces a voltage in secondarywinding 115. When transistor 101 is turned off, a secondary current isinduced in secondary winding 115. The secondary current charges upcapacitor 111 through a diode 110 and provides an output voltage Vout toa load 112. When the energy in the magnetic field is completelydischarged, capacitor 111 maintains an output current flowing in load112 by partly discharging after the secondary current stops flowing.

As shown in FIG. 1, SMPS 100 can include one or more auxiliary windings116, which provides operating power VCC to controller 103 through acircuit that includes resistor 105, diode 106, and capacitor 107.Auxiliary winding 116 also provides a feedback voltage signal Vfb to anFB terminal of controller 103 through resistors 108 and 109. Feedbacksignal Vfb is used by controller 103 as a measure of output voltage Voutof the power supply in order to regulate the output voltage.

When transistor 101 is turned on, primary current Ip through a currentsensing resistor 104, which provides a current sense voltage Vcs to a CSterminal of controller 103. Current sense signal Vcs is used to forcontrolling current flow in the primary winding. For example, transistor101 is turned off, if Vcs reaches a predetermined value.

Even though in wide use, conventional power supplies such as SMPS 100suffer from many limitations. For example, the power transistor employedin the power supply system has a switching frequency limit, in part, dueto the time required to fully turn on or turn off the power transistor.

Another problem can arise when current sense resistor 104 is improperlyselected or malfunctions. As described above, the switch-on time can becontrolled by current sense voltage Vcs. When voltage Vcs reaches apre-determined voltage level, switch 101 is turned off. However, if thecurrent sense resistor is shorted or changed to a lower value, the sensevoltage may not reach the target level. Then the switch control signalmay stay at the on state, and the base drive current would continue toflow through the power transistor. In this case, the primary sidecurrent will continue to increase, and the power transistor or othercomponents maybe damaged due to over heating. The problems describedabove can be further appreciated with the waveforms in FIG. 2.

FIG. 2 is a timing diagram illustrating waveforms of signals inswitching mode power supply system 100 of FIG. 1, and includes thefollowing signals:

-   -   “SW” is a power control signal pulse provided by controller 103;    -   “Base current” is the current provided to the base terminal of        power transistor 101;    -   “Vice” is the collector-emitter voltage of power transistor 101;        and    -   “Ip” is the current flowing through power transistor 101 and        primary winding 114.

As shown by the “Base current” waveform in FIG. 2, at time t1, whenswitch control signal (SW) turns to a high voltage level, a forward basedrive current is applied with an overdrive current followed by aconstant driving current. At time t2, when the primary winding currentIp reaches a pre-determined value, the forward base drive current is cutoff. However, as shown in the “Ip” waveform, primary current Ipcontinues to flow due to stored charges, as indicated by the shaded areaunder waveform Ip. As a result, the response of the power supply to thecontrol signal can be slowed, and the switching frequency can belimited.

Accordingly, there is a need for methods and circuits for moreeffectively controlling the power transistor in a power supply.

Embodiments of the present invention provides a power supply controllerthat includes faster turn on of the power transistor and a dischargecircuit having a low resistance path for discharging the storage chargeof the power transistor. Some examples are described below.

FIG. 3A is schematic/block diagram of switching mode power supply system(SMPS) 300 according to an embodiment of the present invention. Asshown, SMPS 300 includes a transformer 314 that has a primary windingand a secondary winding (not labeled explicitly). SMPS 300 also has apower transistor 301 coupled to the primary winding. Power transistor301 is also coupled to a power supply controller 330, which isconfigured to issue a control signal at its OUT pin to power transistor301. A voltage VCS across a resistor 302 is used to sense a currentflowing through power transistor 301 and the primary winding. SMPSsystem 300 also has other standard SMPS components as shown in SMPS 100of FIG. 1, such as auxiliary winding, feedback circuits, and rectifyingcircuits, etc., which are omitted in FIG. 3A to reduce the complexity ofthe drawing.

As shown in FIG. 3A, power supply controller 330 includes a powercontrol logic 340, which, in some embodiments, may have standard controlfunctions similar to conventional controllers such as controller 103 ofFIG. 1. Depending on the embodiment, power control logic 340 can alsoinclude pulse frequency modulation (PFM) or pulse width modulation (PWM)control logic circuits. Power supply controller 330 also includes adriver circuit 350 which is configured to provide improvement in drivingthe power transistor according to embodiments of the present invention.

Some features of the embodiment of FIG. 3A can be described using thewaveform diagrams in FIG. 3B. For example, in FIG. 3B, waveform “SW”shows the power supply control signal pulse, similar to signal SW inFIG. 2, and waveform “Drive current” in FIG. 3B is the drive signal to acontrol terminal of the power transistor, corresponding to “Basecurrent” in FIG. 2. Additionally, waveform “VCS” in FIG. 3B shows avoltage on the primary current sense resistor (302 in FIG. 3A),corresponding to primary current “Ip” in FIG. 2.

As shown by waveform “Drive current” in FIG. 3B, power supply controller350 is configured to provide a drive current to a control terminal of apower transistor (e.g., the base terminal of power transistor 301) inthree time intervals:

-   -   During a first time interval t1, a first drive current includes        a current spike for turning on the power transistor in response        to a start of the control signal pulse SW;    -   During a second time interval t2, a second drive current        includes a ramping current having a magnitude substantially        proportional to a magnitude of the primary current; and    -   During a third time interval t3, current flow to the power        transistor is at least partially turned off before the end of        control signal pulse SW.

In some embodiment, a method for driving a bipolar power transistorincludes applying driving signals in several stages. During a first timeperiod (including times t1 and t2), the power transistor is turned onwith an initial drive current, which can be started in response to apower supply control signal. An example is shown as the initial step inthe “Drive current” waveform in FIG. 3B. As shown in the “VCS” waveform,which is the voltage on the current sense resistor and reflects themagnitude of the primary current Ip, a current spike is built up.Embodiments of the invention provide a method for increasing the powertransistor drive current with a current related to this leading edgecurrent spike on a current sense resistor during time t1. Then, duringtime t2, current flow in the power transistor is maintained with aramping drive current. When it comes time to turn off the powertransistor, during a second time period (shown as t3), the “Drivecurrent” is turned off, but current flow in the power transistorcontinues to flow, in part due to the stored charges in the powertransistor. This is shown as the rising VCS curve during time t3.

FIG. 3C is circuit schematic diagram of switching mode power supplysystem 300 showing a specific implementation of driver circuit 350 ofFIG. 3A according to an embodiment of the present invention. In FIG. 3C,the controller block 330 of FIG. 3A has been omitted to simplify thedrawings.

Driver circuit 350 in FIG. 3C, can be part of the power supplycontroller chip. Alternatively, it can be implemented in a separateintegrated circuit chip. In an embodiemnt, driver current 350 isconfigured to receive a switch control signal SW from a control logic inthe power supply controller and provide a power transistor controlsignal at an output node OUT, also marked as OUT pin in FIG. 3C. In thisembodiment, driver circuit 350 has a PMOS transistor 303 and an NMOStransistor 308 coupled through the OUT node to power transistor 301. Asdescribed below, transistors 303 and 308 are used to speed up theswitching speed of power transistor 301. In some embodiments, transistor308 is used to speed up the discharge of the power transistor, and isreferred to herein as the discharge transistor or the low-sidetransistor. Driver circuit 350 also includes several functions blocks,such as an internal bias block 311, an overdrive block 312, and apre-shutdown block 313. The overall function of base drive circuit 350and detailed operation of internal bias block 311, overdrive block 312,and pre-shutdown block 313 are described below, with reference towaveforms depicted in FIG. 3D, which is a timing diagram illustratingwaveforms of various signals in the switching mode power supply systemof FIG. 3C according to an embodiment of the present invention.

When switch control signal SW is high, internal bias block 311 providesthe bias for the first overdrive current. As shown in FIG. 3C, internalbias block 311 provides the bias voltages to the gate terminals of NMOStransistors 306 and 307 for the first bias current. Control signalsPRE_SHDN and SW_SHDN, which are provided by pre-shutdown block 313, areset at high levels. Then the first bias current is mirrored by PMOS 304and 303 to generate the first over drive current.

Once power transistor 301 turns on, the OUT pin voltage begins to rampup and is coupled to internal bias block 311 through a feedback path toform a combined bias voltage Vbias. As shown in the waveform diagrams inFIGS. 3B and 3D, a voltage spike in VCS is present on current senseresistor 302, when the power transistor 308 is turned on. In embodimentsof the invention, the voltage spike in VCS is introduced to bias block311 through the Out Pin and the feedback path. This voltage spikeappears as a higher bias voltage Vbias, which is applied to NMOStransistor 306 and 307 and mirrored to PMOS transistor 303, thusproviding a higher drive current spike to switch power transistor 301.

As shown in FIG. 3D, after a pre-determined time interval, as marked bytime t1 associated with waveform “Drive current” in FIG. 3D, thecombined bias voltage Vbias is pulled down to GND by a narrow pulseinterval PPS, and the drive current will be down to a minimum level.Then, during time interval t2, bias voltage Vbias is supplied only bythe OUT pin voltage, so the drive current increases with a slopeproportional to the primary current.

When the current sense voltage VCS reaches a pre-shutdown level,VCS_PRE, the pre_shutdown block 313 provides control signal of PRE_SHDN,which cuts off the major drive current for a time interval, marked by T2in FIG. 3D. The remaining forward drive current will also be cut off.Note in FIG. 3D, signal PRE_SHDN remains high in time period T1 whichincludes the two stages of drive current flow. After time period T2,discharge transistor (low-side transistor) 308 is turned on to speed upthe discharge of the power transistor.

An embodiment of the invention provides a method for turning off abipolar transistor, such as power transistor 301. The method includescoupling an MOS transistor to a base of the bipolar transistor, andbiasing the MOS transistor is bias in two stages. During a first timeperiod, a first voltage is applied to a gate of the MOS transistor todischarge the bipolar transistor. And during a second time period, asecond voltage is applied to the gate of the MOS transistor. The firstvoltage is a higher voltage and overdrives the MOS transistor for fastdischarge of the bipolar transistor. The second voltage is lower thanthe first voltage, but is sufficient to maintain a moderate Rdson.

In this embodiment, the over_drive block 312 provides an overdrivenvoltage VGN to the gate of the low-side transistor 308 in the drivecircuit. As shown by the waveform “VGN” in FIG. 3D, during time periodToff1, low-side transistor 308 will be overdriven to turn on rapidly toprovide a low resistance path to discharge the storage charges to speedup the turn off process of the power transistor. As shown in FIG. 3C,VGN is at GND, when the SW signal is high.

During time period Toff2, when the control signal SWNS becomes highlevel, the gate voltage of 308 will be set to Vdd, an internal powersupply voltage, to ensure the enough gate voltage during the long offtime when the switch control signal SW is low. As shown in FIG. 3C, thecoupling between over_drive block 312 and discharge transistor 308includes control signals SWNPS and SWNS, which are provided by PowerControl Logic 340 in some embodiments. In a specific embodiment, theinternal Vdd may be 5V, which is derived from a power controller chipsupply voltage VCC, which may be more than 10V.

Embodiments of the present invention also provide a method for shortcircuit protection of current sense resistor in the primary side. Asshown in FIG. 3C, when current sense resistor 302 is shorted or changedto a lower value, the power transistor is only turned on by the initialoverdrive current. Virtually no further base drive current is generateddue to the lower voltage on the current sense resistor, so the powertransistor will turn on for a short period of time and then be cut off.Thus the primary current will not continue to increase beyond thecapability to ensure the safety of the power transistor and the wholesystem.

FIG. 4 is a circuit diagram illustrating a specific implementation ofbias block circuit 311 of the switching mode power supply system 300 ofFIG. 3C according to an embodiment of the present invention. As shown inFIG. 4, bias block circuit 400 includes a bipolar transistor 401, aresistor 402, MOS transistors 403 and 406, and a current mirror circuitformed by transistors 404 and 405. Circuit 400 also includes threeswitches controlled by switch signals PS1, PS4, and PPS, respectively.In an embodiment, these switch signals are generated by control logiccircuits in the power controller. Circuit block 400 also receives a biassignal “Internal Bias” and the “OUT Pin” signal, which is coupled to acontrol terminal of the power transistor as described above.

In an embodiment, bias block circuit 400 is configured to generate biasvoltage Vbias, which can provide a higher over current and a ramping updrive current proportional to the current of the primary winding. Whenswitch control signal SW is high, control signal of PS1 and PS4 willturn on the switch controlled by them respectively. The internal biasvoltage in FIG. 4 provides the initial bias voltage. Then the firstover-drive current will be generated, with PMOS transistor 405 suppliesthe overdrive current to the external power transistor because of thevoltage spike on OUT Pin. As a result of the voltage spike, the OUT pinvoltage begins to ramping up and is sent to the internal base terminalof 401. Then internal bias voltage bias2 and the OUT pin voltage turn ontransistors 401 and 403, respectively. The increased current isreflected in he current of PMOS 405 will be increased to provide thehigher over-drive current. As shown, circuit block provides the combinedbias voltage Vbias.

As shown time interval t1 in FIG. 3D, after a pre-determined timeinterval, the combined bias voltage Vbias is pull down to GND by narrowpulse signal PPS, then the drive current will decrease to a minimumvalue. In FIG. 4, the switch controlled by control signal PS1 will beturned off, and the combined bias voltage Vbias will be only supplied byOUT pin voltage. Subsequently, during time interval t2 in FIG. 3D, thedrive current will increase with a slope proportional to the primarywinding current. In an embodiment, the selection of the pre-determinedtime interval for resetting combined bias voltage Vbias is intended toprovide sufficient drive current to the power transistor, but withoutoverly charging it, because excess storage charges tend to prolong thedischarging time and limit the useful frequency range of the powersupply.

The embodiments describe above can also provide protection functionsagainst certain malfunctions of the power supply control circuit. Forexample, in the event that the current sense resistor is shorted orchanged to a lower value, the OUT pin voltage is drop down also and cannot be added for the combined voltage. So the bias block circuit onlyprovides a initial bias voltage “Internal bias” to generate the initialoverdrive current, virtually no further base drive current is generated,so the power transistor will turn on for only a short period of time andthen be cut off. The primary current will not continue to increasebeyond its capability to ensure the safety of the power transistor andthe whole system.

FIG. 5A is a circuit diagram illustrating a specific implementation ofthe pre-shutdown 313 circuit of the switching mode power supply system300 of FIG. 3C according to an embodiment of the present invention. FIG.5B is a timing diagram illustrating waveforms of various signals in thepre-shutdown circuit 600 of FIG. 5A according to an embodiment of thepresent invention. As shown in FIG. 5A, pre-shutdown block 600 includesa capacitor 601, a multiplexer circuit 602, a comparator 603, a latchcircuit formed with gates 604 and 605, and additional logic gates.Current sources I1 and 12 and switches SWN, P1, and P2 are configuredfor charging and discharging of capacitor 601.

As shown in FIG. 5B, when switch control signal SW is high, powertransistor 308 is turned on, and the current sense voltage VCS begins toincrease. The switch control signal P1 is high and the capacitor 601 isbeing charged with the initial value of a fixed voltage VCS_PRE. Thepositive terminal of comparator 603 is connected with the outputterminal of the 2 to 1 multiplexer 602, and the negative terminal ofcomparator 603 is connected to the internal pre-shutdown level VCS_PRE.One input terminal of multiplexer 602 is connected with VCS, the currentsense voltage associated with the primary current. The other terminal ofthe 2 to 1 multiplexer 602 is connected with capacitor 601 to received avoltage VC1. The enable signal “M” of the multiplexer 602 will selectwhich input terminal is sent to the input of the comparator 603.

In FIG. 3D, the waveform for “PRE_SHDN” includes two time periods, T1and T2. During T1, a drive current is provided to the power transistor,as shown during t1 and t2 of the “Drive current” waveform. During timeperiod T2, the drive current is cut off. The ratio of T1 to T2 isrelated to the ratio of constant current sources 1 and 2 in FIG. 5A. Forexample, in a specific embodiment, I2=4*I1, and therefore, T2 is 25% ofT1. In an embodiment, the duration of T2 is selected according to acharge storage time of power transistor 301 in FIG. 3C.

FIG. 5B also shows the waveform for “PRE_SHDN” including two timeperiods, T1 and T2. As shown in FIG. 5B, when current sense voltage VCSor CS_A reaches to the internal pre-shutdown level VCS_PRE, the outputof the comparator 603 will change to a high level, then a switch controlsignal P2 is high to discharge the capacitor 601. The enable signal “M”of multiplexer 602 sets the capacitor voltage VC1 connected to positiveinput terminal of the comparator 603. When VC1 dropped to pre-shutdownlevel VCS_PRE, the switch control signal SW is turned off, and theenable signal “M” of multiplexer 602 causes the current sense value VCSto be connected to positive input terminal of the comparator 603 again.The capacitor voltage VC1 will be reset with the fixed voltage ofVCS_PRE by control signal SWN.

In this embodiment, T2 is the duration of pre-shutdown control signalP2, and determines the time for cut off the major base drive current. Bysetting the ratio of charging current I1 and discharging current 12,time T2 can be selected, as described above.

FIG. 6 is a circuit diagram illustrating a specific implementation ofthe overdrive block circuit 312 the switching mode power supply systemof FIG. 3C according to an embodiment of the present invention. In anembodiment, overdrive block 500 provides an overdrive signal VGN to thedischarge transistor or low-side transistor 308. As shown, overdrivecircuit 500 includes a current source which, in this embodiment, isconfigured as a 1:N1 current mirror circuit formed with MOS transistors504 and 505, where the current ratio N1 can be selected depending on theembodiment. Overdrive circuit 500 has a first circuit that includes MOStransistor 503, resistor 502, and capacitor 501. The first circuit iscoupled to the current mirror through a switch controlled by signalSWNPS. The first circuit also receives a bias signal labeled InternalBias. Overdrive circuit 500 also has a second circuit that includes abipolar transistor 511, a resistor 510, and one or more diode devices506-509 connected in series.

When the power controller determines that the power transistor needs tobe turned off, control signal SWNPS is used to turn on the switchcoupled to the current mirror to generate current in NMOS 503. Thecurrent is then mirrored to PMOS 505. As shown in FIG. 5, output signalVGN is set by the forward voltages of the series diodes 506-509 and aforward base-emitter diode of NPN transistor 511. As shown in FIG. 3,the VGN signal is connected to the gate of low-side transistor 308 ofbase drive circuit 350 when SWNPS is high.

In some embodiments, e.g., as shown in FIG. 6, the current mirror iscoupled to power supply VCC of the power supply controller, which ishigher than the internal power supply Vdd inside the power supplycontroller. As shown in FIGS. 3B and 6, the gate of low-side transistor308 can be coupled to either VCC or Vdd, depending on the switchescontrolled by signals SWNS and SWNPS. For example, in some embodiments,VCC can be more than 10V, whereas Vdd is an internal bias voltage derivefrom VCC and may be 5V. Of course, different VCC and Vdd can be selectedfor different applications. In some embodiments, with overdrive block500, VGN can be larger than the internal power supply Vdd, so low-sidetransistor 308 is overdriven in the first stage of turning off the powertransistor, as shown in FIG. 3D.

As shown in FIGS. 3C and 3D, when the power supply controller turnscontrol signal SWNS to a high level, the gate voltage of 308 is set tointernal power supply voltage Vdd, to maintain the gate voltage duringthe power transistor 301 off time when the switch control signal SW islow.

As described above with reference to FIG. 3D, when switch control signalSW is high, the power transistor is turned on by an initial overdrivecurrent. Then the voltage on the primary current sense resistor VCS issent to internal bias block 311, which with an internal bias voltageoutputs a higher combined bias voltage Vbias to provide a higheroverdrive current.

After a pre-determined time interval, the combined bias voltage Vbiaswill be pull down to GND by a narrow pulse control signal PPS and thedrive current decreases to a minimum value. Then the bias voltage Vbiaswill be only supplied by the OUT pin voltage, so the drive currentincreases with a slope proportional to the primary winding current. WhenVCS voltage reaches to its internal pre-shutdown voltage VCS_PRE, themajor base drive current will be cut off for a time interval that is afraction of the power switch turn on time. The remaining forward drivecurrent will also be cut off and low-side transistor 308 will be turnedon.

As described above, signal SWNPS controls the gate overdrive voltagecircuit to provide a overdriven voltage VGN, which is applied to thegate of the low-side transistor of the base drive circuit. In someembodiment, VGN can be larger than the internal power supply Vdd, so thelow-side transistor 308 is overdriven in the first stage to turn off thepower transistor. When the control signal SWNS becomes high level, thegate voltage of the low-side transistor 308 will be Vdd, the internalpower supply voltage to ensure the enough gate voltage during the powertransistor off time when the switch control signal SW is low.

While the preferred embodiments of the invention have been illustratedand described, it will be clear that the invention is not limited tothese embodiments only. Numerous modifications, changes, variations,substitutions and equivalents will be apparent to those skilled in theart without departing from the spirit and scope of the invention.

What is claimed is:
 1. A driver circuit for turning off a bipolartransistor, the driver circuit comprising: a current mirror having firstand second PMOS transistors coupled to a first voltage during a firsttime period; a bipolar transistor coupled to the current mirror; aplurality of diode devices coupled between a collector and a base of thebipolar transistor; and an MOS transistor having a gate terminal coupledto the collector of the bipolar transistor and a drain terminal forcoupling to a control terminal of the bipolar transistor for turning offthe power transistor.
 2. The driver circuit of claim 1, wherein the gateof the MOS transistor is configured to be coupled to a second voltageduring a second time period, the second voltage being lower than thefirst voltage.
 3. The driver circuit of claim 1, further comprising: aninput terminal coupled to the current mirror for receiving a timingsignal; a second MOS transistor coupled between the input terminal and aground terminal; and a resistor and a capacitor coupled in seriesbetween a gate of the MOS transistor and the ground terminal.
 4. Amethod for turning off a bipolar transistor, comprising: coupling an MOStransistor to a base of the bipolar transistor; during a first timeperiod, applying a first voltage to a gate of the MOS transistor todischarge the bipolar transistor; and during a second time period,applying a second voltage to the gate of the MOS transistor, the secondvoltage being lower than the first voltage.
 5. The method of claim 4,further comprising at least partially reducing a current flow to thebase of the bipolar transistor before the first time period.
 6. Themethod of claim 4, further comprising deriving the first voltage from aplurality of serially connected diode devices and a base-emitter voltageof a second bipolar transistor.